Frequency drift estimation for low cost outdoor unit frequency conversions and system diagnostics

ABSTRACT

Systems and devices for controlling frequency drift in satellite broadcast systems. A receiver antenna system for a direct broadcast satellite signal communications system in accordance with one or more embodiments of the present invention comprises an oscillator, a mixer, coupled to the oscillator, for converting satellite signals at a first frequency to signals at an intermediate frequency, an analog-to-digital (A/D) converter, coupled to the mixer, for receiving the signals at the intermediate frequency and for converting the signals at the intermediate frequency at near-real-time to a digital data stream, a Digital Signal Processor (DSP), coupled to the A/D converter, for processing the digital data stream, and a drift estimator, coupled to the DSP, the drift estimator determining a frequency drift of the oscillator, wherein the receiver antenna system corrects the frequency drift of the oscillator using the determined frequency drift.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is related to U.S. application Ser. No. 12/195,256,filed on Aug. 20, 2008, by Robert F. Popoli, entitled “COMPUTATIONALLYEFFICIENT DESIGN FOR BROADCAST SATELLITE SINGLE WIRE AND/OR DIRECT DEMODINTERFACE,” which application is incorporated by reference herein.

This application claims the benefit under 35 U.S.C. Section 119(e) ofU.S. Provisional Application Ser. No. 61/142,865, filed on Jan. 6, 2009,by Robert F. Popoli, entitled “FREQUENCY DRIFT ESTIMATION FOR LOW COSTOUTDOOR UNIT FREQUENCY CONVERSIONS AND SYSTEM DIAGNOSTICS,” whichapplication is incorporated by reference herein.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates generally to satellite video systems, andin particular, to a method, apparatus, and article of manufacture forfrequency drift estimation in satellite outdoor unit systems.

2. Description of the Related Art

Satellite broadcasting of communications signals has become commonplace.Satellite distribution of commercial signals for use in televisionprogramming currently utilizes multiple feedhorns on a single OutdoorUnit (ODU) which supply signals to up to eight IntegratedReceiver/Decoders (IRDs), e.g., set top boxes, on separate cables from amultiswitch.

FIG. 1 illustrates a typical satellite-based broadcast system of therelated art.

System 100 uses signals sent from Satellite A (SatA) 102, Satellite B(SatB) 104, and Satellite C (SatC) 106 that are directly broadcast to anOutdoor Unit (ODU) 108 that is typically attached to the outside of ahouse 110. ODU 108 receives these signals and sends the received signalsto IRD 112, which decodes the signals and separates the signals intoviewer channels, which are then passed to monitor 114 for viewing by auser. There can be more than one satellite transmitting from eachorbital location (slot). The orbital slots are typically designated bytheir longitude, so, for example, a satellite 102 located in the orbitalslot at 101 degrees West Longitude (WL) is usually referred to astransmitting from “101.”

Satellite uplink signals 116 are transmitted by one or more uplinkfacilities 118 to the satellites 102-106 that are typically ingeosynchronous orbit. Satellites 102-106 amplify and rebroadcast theuplink signals 116, through transponders located on the satellite, asdownlink signals 120. Depending on the satellite 102-106 antennapattern, the downlink signals 120 are directed towards geographic areasfor reception by the ODU 108.

Each satellite 102-106 broadcasts downlink signals 120 in typicallythirty-two (32) different frequencies, which are licensed to varioususers for broadcasting of programming, which can be audio, video, ordata signals, or any combination. These signals are typically located inthe Ku-band of frequencies, i.e., 11-18 GHz, but can also be broadcastin the Ka-band of frequencies, i.e., 18-40 GHz, but typically 20-30 GHz.

Within ODU 108, the downlink signals 120 are downconverted to lowerfrequencies using an oscillator and a mixer. Typically, the oscillatoris a Dielectric Resonant Oscillator (DRO). If the DRO frequency drifts,the downconversion of the downlink signals 120 drifts as well, whichmakes processing of such downconverted signals more difficult.

Further, as satellites 102-106 broadcast additional services andadditional channels to viewers, as well as additional satellite signalspresent in such bandwidths, it will be more efficient to monitor and, ifnecessary, correct drifts in the DRO frequency.

SUMMARY OF THE INVENTION

To minimize the limitations in the prior art, and to minimize otherlimitations that will become apparent upon reading and understanding thepresent specification, systems and devices for controlling frequencydrift in satellite broadcast systems are presented herein.

A receiver antenna system for a direct broadcast satellite signalcommunications system in accordance with one or more embodiments of thepresent invention comprises an oscillator, a mixer, coupled to theoscillator, for converting satellite signals at a first frequency tosignals at an intermediate frequency, an analog-to-digital (A/D)converter, coupled to the mixer, for receiving the signals at theintermediate frequency and for converting the signals at theintermediate frequency at near-real-time to a digital data stream, aDigital Signal Processor (DSP), coupled to the A/D converter, forprocessing the digital data stream, and a drift estimator, coupled tothe DSP, the drift estimator determining a frequency drift of theoscillator, wherein the receiver antenna system corrects the frequencydrift of the oscillator using the determined frequency drift.

Such a system further optionally comprises the drift estimator driving adigital mixer within the DSP to compensate for the determined frequencydrift, an output of the drift estimator being fed back to the oscillatorto control the frequency drift of the oscillator, an automatic gaincontrol coupled between the mixer and the A/D converter, the A/Dconverter sampling the signals at the intermediate frequency at a speedgreater than 1 gigasample per second, the satellite signals beingtransmitted in at least the Ku-band of frequencies, the satellitesignals being further transmitted in at least the Ka-band offrequencies, a Digital-to Analog Converter (DAC), coupled to the DSP,and an output of the DAC being mixed with a second oscillator, such thatan output of the receiver antenna system is set to a desired band on asingle wire interface.

A system for distributing a plurality of satellite signals on a singleinterface in accordance with one or more embodiments of the presentinvention comprises an oscillator for down-converting the plurality ofsatellite signals to signals at an intermediate frequency, an AutomaticGain Controller (AGC) for gain controlling the signals at theintermediate frequency, an analog-to-digital (A/D) converter, coupled tothe AGC, for receiving the signals at the intermediate frequency,wherein the A/D converter directly samples the signals at theintermediate frequency, and a Digital Signal Processor (DSP), coupled tothe A/D converter, wherein a first output of the DSP is used todetermine the intermediate frequency and a second output of the DSP isan input to the single interface.

Such a system further optionally comprises a Digital-to-Analog Converter(DAC), coupled to the second output of the DSP, the first output of theDSP determining the intermediate frequency by controlling a frequency ofthe oscillator, the first output of the DSP driving a compensatoryfrequency shift by controlling a digital mixer internal to the DSP, theoscillator being a Dielectric Resonance Oscillator (DRO), the pluralityof satellite signals being transmitted in a plurality of frequencybands, the plurality of frequency bands comprising a Ka-band and aKu-band, the DRO down-converting the Ka-band to at least a firstintermediate frequency band and the DRO down-converting the Ku-band to asecond intermediate frequency band as convenient for A/D conversion, theA/D converter sampling the signals at the intermediate frequency at arate greater than the Nyquist rate for the signals at the intermediatefrequency, state information from at least one of the AGC, the ADC, theDSP, and the DAC providing a diagnostic output for the system, and thediagnostic output comprising at least one of fault recognition, faultreporting, performance monitoring, installation aiding, and installationverification for the system.

Other features and advantages are inherent in the system disclosed orwill become apparent to those skilled in the art from the followingdetailed description and its accompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

Referring now to the drawings in which like reference numbers representcorresponding parts throughout:

FIG. 1 illustrates a typical satellite-based broadcast system of therelated art;

FIG. 2 illustrates a typical Single Wire Multiswitch (SWM) of therelated art;

FIGS. 3A and 3B illustrate related LNB modules and an LNB module inaccordance with one or more embodiments of the present invention,respectively;

FIG. 4 illustrates an Analog-to-Digital subsystem functionality inaccordance with one or more embodiments of the present invention;

FIG. 5 illustrates a block diagram of the Digital Single WireMultiswitch (DSWM) Channelizer in accordance with one or moreembodiments of the present invention;

FIG. 6 illustrates a coarse granularity channelizer in accordance withone or more embodiments of the present invention;

FIG. 7 illustrates an exemplary fine granularity channelizer embodimentin accordance with one or more embodiments of the present invention; and

FIG. 8 illustrates a block diagram of a frequency drift control systemin accordance with one or more embodiments of the present invention.

DETAILED DESCRIPTION

In the following description, reference is made to the accompanyingdrawings which form a part hereof, and in which are shown, by way ofillustration, several embodiments of the present invention. It isunderstood that other embodiments may be utilized and structural changesmay be made without departing from the scope of the present invention.

Overview

Recent advances in high speed Analog to Digital (A/D) converters open upthe possibility of direct A/D conversion of baseband or near baseband500 MHz wide Satellite downlink signals. This allows for all-digitaldemultiplexing (demuxing) of one or more 500 MHz complexes ortransponders and subsequent all-digital multiplexing (muxing) of aselected subset of these transponder channels onto a single wireinterface for home distribution. With such a digital implementation,selected transponder baseband I/Q signals may be selected for directdemodulation, for subsequent home or multi-dwelling distribution overany suitable physical/network layer protocols.

By integrating the high-speed A/D hardware withcomputationally-efficient Digital Signal Processing (DSP) techniques,and adapting them specifically for the purposes of demuxing, muxing, andbaseband I/Q conversion, one or more embodiments of the presentinvention provide significant commercial advantages over current analogdesigns. Further, simple brute force digital mimicry of known analogbuilding blocks for demuxing/muxing/conversion processes are expected tolead to less commercially-viable designs.

No known previously proposed satellite home/MDU distribution designexists to create a practical all digital implementation of the demuxing,muxing, and direct digital demodulation (demod) interface. Thisspecification proposes a number of embodiments of such designs whichutilize computationally efficient techniques which allow forcommercially viable all digital implementation of these functions.

The present invention allows for better overall performance of anembodiment of a Single Wire Multiswitch (SWM) architecture because thepresent invention allows for the distribution of more channels withinthe same bandwidth (i.e., single wire bandwidth) through tighter packingof channels. Distribution of additional channels allows for the supportof additional IRDs, or the same number of IRDs with less wiredbandwidth. Further, embodiments of the present invention allow forinexpensive provision of baseband I/Q signals for reduced costintegration of a significant portion of the current IRD. Further still,embodiments of the present invention allow for the ability to providegreater flexibility for future system designs, and the potential of asmaller footprint for the SWM into the current system. Embodiments ofthe present invention also allow for simpler integration and interfacesbetween the parts of the present system, by simplifying the interface ofshared demodulation resources without expending any bandwidth. Thissimplification allows for future products, such as Home Gateway andMulti-Dwelling Unit (MDU) architectures, to become possible.

Digital SWM (DSWM)

FIG. 2 illustrates a typical Analog Single Wire Multiswitch.

Hardware advances have increased A/D sampling rates in excess of 1Gigasample/second (1 GSPS) with a good Effective Number Of Bits (ENOB)and an adequate linearity performance figure. Advances in multi-rateDigital Signal Processing techniques coupled with nanometerApplication-Specific Integrated Circuit (ASIC) processes allowapplications with significant signal processing capabilities. Thesefactors make it possible to make an all-digital replacement for thedemux and mux functions of the Single Wire Multiswitch (SWM). Further,these technological advances allow for a baseband and/or near basebanddigital I/Q interface for cost-effective integration of a significantportion of the IRD front end functionality.

One or more embodiments of the present invention provide a digitalreplacement for the hardware shown in FIG. 2. FIG. 2 illustrates SWM200, where SWM 200 is fed by four composite signals 202-208,specifically LNB1 202, LNB2 204, LNB3 206, and LNB4 208, which areproduced by the LNB module 300 hardware shown in FIG. 3A.

Each LNB signal 202-208, as shown in FIG. 3A, comprises multiple (e.g.,three) stacked 500 MHz bandwidth signals, generated from downlinksignals 120 and received at various orbital slots. Ku-band signals 302are from satellites at the 119 WL slot, Ku-band signals 304 are fromsatellites at the 110 WL slot, Ka-band signals 306 are from satellitesat the 102.8 (also referred to as 103) WL slot, Ku-band signals 308 arefrom satellites at the 101 WL slot, and Ka-band signals 302 are fromsatellites at the 99.2 (also referred to as 99) WL slot. Combinations ofthese signals, based on their polarization and transmission frequencies,are used to generate LNB signals 202-208.

After down-conversion within module 300, the Ku-band signals 302, 304,and 308 are down-converted to an Intermediate Frequency (IF) band, 500MHz wide, in the frequency range of 950-1450 MHz. The Ka-band signals306 and 310 are down-converted into two different 500 MHz widebandwidths, namely the 250-750 MHz bandwidth (known as Ka-LO IF band orKa-B IF band) and the 1650-2150 MHz bandwidth (known as Ka-HI IF band orKa-A IF band), and are combined in various combinations to form signals202-208

The SWM 200 shown in FIG. 2 selects any nine 40 MHz pieces of spectrumfrom LNB1 202 through LNB4 208 and stacks them to form a singlecomposite signal called the channel stacked output 210. The nine 40 MHzchannels are typically located on 102 MHz centers and range from 974 MHzto 1790 MHz (i.e., the channels are at 974, 1076, 1178, 1280, 1382,1484, 1586, 1688, and 1790 MHz, respectively).

Coarse Granularity Design

FIG. 3B illustrates an embodiment an LNB structure corresponding to aDigital SWM Design in accordance with the present invention.

The SWM module 300 is modified into module 312, where signals 302-310are combined into different signals that are to be used as inputs to amodified SWM. Rather than stacking 1500 MHz into a single signal, as isdone with signals 202-208, a larger number of outputs 312-326 are used.Although eight outputs 312-326 are shown, a larger or smaller number ofoutputs 312-326 are possible without departing from the scope of thepresent invention.

As shown in FIG. 3B, LNB1 signal 312, LNB2 signal 314, LNB5 signal 320,and LNB6 signal 322 each comprise two 500 MHz signals, each 500 MHzsignal corresponding to a Ka-LO band signal and a Ka-HI band signal.This arrangement is appropriate if the A/D has enough front endbandwidth to subsample the Ka-Hi band in its 2^(nd) Nyquist Band.Alternatively, separate independent IF conversion for the Ka-Lo andKa-Hi can be provided for more traditional 1^(st) Nyquist band samplingof each signal. LNB3 signal 316, LNB4 signal 318, LNB7 signal 324, andLNB8 signal 326, on the other hand, are 500 MHz signals, eachcorresponding to a Ku-band signal. To facilitate A/D conversion, thelocal oscillators 328 can be modified so that each of the signals312-326 can have a desired and, possibly, different tunable startingfrequency from 10 to 100 MHz, or beyond these limits if desired.Additional mixing can be added to achieve an offset frequency start forone or more of the signals 312-326 if desired. It is also possiblewithin the scope of the present invention to implement tuning forsignals 312-326 within the digital domain if desired.

FIG. 4 illustrates an embodiment of the Analog-to-Digital subsystemfunctionality of the present invention.

FIG. 4 shows the A/D subsystem 400 functions, where the samplingfrequency Fs is adjustable, typically from 1.02 GHz to 1.2 GHz, butother frequencies and ranges are possible within the scope of thepresent invention. Each signal 312-326 is placed through a filter 402,and then into an A/D converter 404, which produce the output signals406-428. The typical maximal sampling rate of the A/D converters 404,FsMAX, is typically 1.35 GHz, but other rates are possible withoutdeparting from the scope of the present invention. The Ka HI signals aretypically sub-sampled, so the analog paths should have a useablebandwidth of FsMAX+500 MHz, which is typically 1.85 GHz. Alternativelyif separate IF conversion is provided for the Ka-Hi signal thentraditional 1^(st) Nyquist band sampling and Lo-Pass anti-alias filterswould be employed.

FIG. 5 illustrates a block diagram of the DSWM Channelizer in accordancewith one or more embodiments of the present invention.

Each input “x” 500, 502, etc., receives one of the A/D converter 404outputs 406-428, and there can be extra inputs x 500, 502, etc., toallow for expansion of the system. Each typical input x 500 is typicallychannelized into uniformly spaced K filters 504. The portion k 506 ofthe K filters 504 actually utilized may be different for different LNBpaths, and, typically, k=12 for Ka HI, k=12 for Ka LO, and k=16 for Ku.L, the number of filter bands on the output side, is set to be equal tothe number of stacked carriers desired at the stacked output 210.

There are many embodiments within the scope of the present inventionthat can create computationally efficient DSP architectures. Anotherexample embodiment of the present invention uses multi-rate poly phasetechniques and takes advantage of the correspondence between the complexmixing and the Fast Fourier Transform (FFT), as shown in FIG. 6.

FIG. 6 illustrates a coarse granularity channelizer in accordance withone or more embodiments of the present invention.

In FIG. 6, system 600 shows inputs 406-428, each entering a Real toComplex Hilbert transformer 600. Other types of Real to Complextransforms of the signal inputs 406-428 are also possible within thescope of the present invention. The K filters 604 for each of the inputs406-428 are set to 16, but other settings are possible within the scopeof the present invention, including different values for K and x foreach input 406-428. If the transponder count and spacing are uniform andwell known, then K can be set to the number of transponders, etc. If,however, the number of transponders varies from satellite to satelliteand/or the spacing of these transponders is not uniform, then a numberof filters 604 greater than the maximum number of transponders to beencountered is typically used to allow for expansion. Further, thefilters are then expanded or contracted by slight adjustments in thesampling frequency and by slight shifts in the down-converter LOfrequency used in the LNB. By adjustment of both the down-converter LOof the LNB and sampling rate, any channel spacings can be accommodatedin any of the LNB outputs.

The signals 406-428, after being filtered by filters 602 are subjectedto Fast Fourier Transforms (FFT) in FFTs 606, and then selected,reordered, and combined in multiplexer 608. The combined signal isfurther processed to generate stacked output 210. Alternatively, thepresent invention can use a non-maximally decimated filter bank ofoverlapping filters. This approach, along with the added technique ofnear-perfect reconstruction techniques simplify the fine granularitydesign.

A choice of L=32 and K=16 are illustrative only. The choice of the powerof 2 composite K and L is chosen to simplify the FFT hardwareimplementation. Other approaches or arrangements may also be used withinthe scope of the present invention.

ADC ENOB Considerations

Consider any one of the 500 MHz LNB signals 406-428. Since each signalis a composite of many signals, each signal is approximately Gaussiandistributed. Given this approximation, the attack point on the ADC canbe set low enough so that probability of the input exceeding the fullscale deflection is controlled and thereby minimize the clipping noiseeffects. Similarly the attack point can be set high enough so that theADC is driven hard enough to minimize the effects of quantization noise.The balancing of these two noise affects yields an optimal attack point(sometimes referred to as backoff point). For any given backoff pointthen the ADC Effective Number of Bits (ENOB) will determine the Signalto Quantization Noise Floor. For anticipated transponder loadings anENOB of 8 bits is more than sufficient. Since A/D are available atapproximately this ENOB while operating well in excess of 1 Gsps, thecurrent state of the art in ADCs is sufficient to implement the designsdescribed by this invention. However, an accounting of the Noise-PowerRatio of the finite word processing and additional details of the noisebandwidth and system settings may allow the ENOB to be relaxed from a 8bit setting and still be within the scope of the present invention.

Fine Granularity Design

FIG. 7 illustrates a fine granularity channelizer implementation inaccordance with one or more embodiments of the present invention.

A fine granularity design using perfect or near perfect reconstructionpolyphase filtering techniques has advantages over the coarsegranularity approach. For such an approach, the spectrum of each of theincoming 500 MHz blocks is subdivided much more finely than in thecoarse granularity design. Where in the coarse granularity design thegoal is to create a number of filters greater than or equal to themaximum number of expected transponders, in the fine granularity designthe goal is to divide up the spectrum into smaller pieces. This can bedone in such a way that the fine pieces of spectrum can be “glued” backtogether to yield a nearly perfect reconstruction of any arbitraryspectral bandwidth within the granularity specified for the design.Employment of non-maximal decimation techniques can be used to simplifyfilter design if desired. An illustrative design for a fine granularitysystem is shown in FIG. 7.

The system 700 shown in FIG. 7 has four output signals 702, 704, 706,and 708. Outputs 702 and 704 are typically used to create two 500 MHzblocks of single wire bandwidth. Outputs 702 and 704 can then be powercombined onto a single wire interface and thus replicate the output ofrelated SWM designs, except that they provide more channels within thesame physical bandwidth.

The other outputs 706 and 708 depicts two illustrative embodiments thatinclude shared demod assets that do not expend any of the Single WireBandwidth. Other embodiments are possible within the scope of thepresent invention. Optionally, only one output 706 or 708 can beimplemented, or other embodiments can be implemented alone or in anycombination, without departing from the scope of the present invention.Output 706 illustrates an approach having an additional internal singlewire interface which drives conventional receiver/demod inputs.

Output 708 illustrates an approach where the receiver portion of thereceiver demod chips can be eliminated at the same time as the lastup-conversion of the processed signals. These chips correspond to I/Qnear baseband demodulation. Other configurations are possible if some ofthe output polyphase filtering is incorporated directly on the demodchip for individual true baseband I/Q processing. In both outputs 706and 708, the output of the shared demod resources is SatelliteCommunicator Identification Code (SCID) filtered data which is networkedonto any suitable physical layer and/or network layer protocols fordistribution throughout the house or Multiple Dwelling Unit (MDU). Theoutputs 706 and 708 are Internet Protocol (IP) type outputs, or similar,that can be output over ethernet cabling, local area networks, RF, orother similar interfaces as desired, without departing from the scope ofthe present invention.

DRO Frequency Drift Control

FIG. 8 illustrates a functional block diagram of processing of one 500MHz band in accordance with one or more embodiments of the presentinvention.

Advances in Digital Signal Processing (DSP) techniques, specifically,the speed at which DSP techniques can now take place, allow for moreefficient bandwidth packing of signals on a single wire interfacebetween ODU 108 and receiver 112. However, in order to take advantage ofthis tighter signal packing, the frequency stability of frequencysources, in this case, the frequency source used to initiallydown-convert signals 120 into an intermediate frequency, must besimilarly improved. The present invention allows for estimation offrequency drift of the frequency source used for down-conversion, aswell as allowing for the ability to measure frequency domain errorswhich will allow for diagnostic feedback on the state of system 100.

As shown in FIG. 8, signals 120 are received at ODU 108, and then arepassed on to receiver 112. Within ODU 108, an antenna 800 receives thesignals 120 at both Ku-band and Ka-band (e.g., approximately 20 GHz),which are mixed at mixer 802 with the output of DRO 804. This mixingprocess results in an output of mixer 802 of the sum and differencefrequencies of signals 120 and DRO 804, and the frequency of DRO 804 ischosen to down-convert signals 120 to a frequency convenient for A/Dconversion.

These signals are then typically filtered through a Band Pass Filter(BPF) 808 to remove harmonics and other unwanted signals, and thentypically passed to Automatic Gain Control (AGC) 810 circuitry tonormalize the signals 806 to a common signal strength. Such a commonstrength is typically desirable to properly excite Analog-to-Digital(A/D) converter 812 to set the attack point as previously described inthe ENOB Considerations discussion.

A/D 812 is a high-speed A/D converter, in that it can run at 1Gigasample per second (GSPS) or higher. Such speed is needed because thetypical bandwidth of the signals 806 is 500 MHz, and to sample at theminimum Nyquist rate for such signals requires a minimum of twice thebandwidth, or 1 GSPS. A/D 812, thanks to recent advances in A/D design,can now achieve 1 GSPS rates, and higher. Once digitized by A/D 812,Digital Signal Processor (DSP) 814 processes the signals and convertsthem back to analog signals in the Digital-to-Analog Converter (DAC)816. Such signals are then mixed at mixer 818 with a local oscillator820 and forwarded on to receiver 112. Optionally, this last mixer can beavoided if a DAC with sufficient BW is employed such that it candirectly cover the desired single wire interface band.

Within DSP 814, the signals can be sent to a drift estimator 822, wheredrift of the frequency of signals 806 can be sensed. Sensing drifts fromvarious sources are achievable within the scope of the presentinvention, e.g., drifts which affect single carriers within a givenbandwidth of signals or drifts which affect an aggregate of carriers,etc., however, the primary cause of drift of signals 806 is the DRO 804frequency drift with temperature and age. DRO 804 is located at the dishantenna 800 of ODU 108, and as the sun heats the dish antenna 800, DRO804 warms up which causes drift, and as the sun goes down dish antenna800 cools down, cooling down DRO 804, again causing drift in frequency.Further, aging of the DRO 804 will also cause a frequency drift in theDRO 804 output.

The drift estimator 822 can be of several varieties. One solution thatcan be used for drift estimator 822 is to create a discriminator basedon a matched filter design, such that the expected type and distributionof transponders within the aggregate bandwidth of signals 806 is sensed.Such a discriminator can be developed for individual transponders ifdesired, or the edges of the bandwidth can be sensed. For transponderswith symmetrical properties, the discriminator can be a matched filterwith a sign reversal about the center frequency, however, if the purposeis to drive the average drift of the DRO 804 to zero, a matched filterfor all of the transponders within the aggregate bandwidth can beutilized. Although the drift estimator algorithm may result in a biasedestimate of each transponder's frequency that is down-converted by theDRO 804, the DRO 804 is tuned by the aggregate of the transponder driftdiscriminants (biased or not) and will therefore tend to average out thebiasing effects due to any individual transponder gain slopes andripples.

The present invention also allows for system 100, via ODU 108 (alsocalled a Single-Wire Multiswitch ODU or SWM-ODU) to recognize largefrequency deviations and frequency trends over the life of ODU 108. AsDRO 804 frequencies drift over the life of DRO 804, the presentinvention can provide not only feedback to DRO 804, but to systemproviders via receiver 112 callbacks or other communications betweenreceiver 112 and system providers. Receivers 112 typically comprise ahigh-speed internet or other interface to allow communications betweenthe system provider and individual receivers 112; reporting system 100health issues can now, through the use of the present invention, beundertaken by processors in receiver 112 and/or DSP 814.

The outputs of drift estimator 822 are then converted to analog voltagesin DAC 824, and then fed back to DRO 804 via a control input 826 to DRO804. In essence, DRO 804 acts like a Voltage Controlled Oscillator (VCO)at this point, where the voltage applied at 826 controls the frequencyoutput of DRO 804.

Although the use of DSP outputs to sense and correct frequency driftfaults is emphasized herein, other uses of DSP 814 outputs to recognizeother faults and to take other corrective actions are possible withinthe scope of the present invention. In particular, by virtue of thearchitecture of the described processing key diagnostic information isavailable for other diagnostics. A mixed signal ASIC will typicallyinclude the AGCs 810, ADCs 812, DSP 814, DACs 816, and a general purposeprocessor implementing the Drift Estimator 822 matched filterdiscriminant. Thus the general purpose processor has access toinformation regarding the state of each of these subsystems. Beyonddetermining the frequency drift, the matched filter results coupled withknowledge of the AGC states provides important diagnostic information ifno match is found over the possible range of drift frequencies of agiven expected transponder. The system can thereby recognize the absenceof expected transponders and can report this fault. This faultrepresents a fault in system 100 to deliver the missing transponder tothe ODU 108. The failure of the match filter to find a match for anytransponder in a given 500 MHz band indicates the potential failure ofone of the 500 MHz signal paths 312,314,318,320, 322,324, and 326 orsubsequent processing these signal paths indicating a potential hardwarefailure in the ODU 108. In particular, if it is known that other ODUs108 at other sites (homes) in the near vicinity of a given ODU 108 arenot reporting faults, then it can be concluded by system 100 that thefault is isolated to the given ODU (“lone ODU”). In contrast, if allODUs 108 in the vicinity of a given ODU 108 all report faults on many500 MHz bands over a given short period of time followed by a completerecovery, the likely cause of the event is a weather generated event. Inparticular, if a collocated group of ODUs report faults occurringprimarily or for longer periods of time on the Ka bands as opposed tothe Ku bands then a weather event is very likely since weather affectsKa signals much more severely then the Ku signals. If a lone ODU 108reports failures on many of its signal paths for extended periods oftime then ODU 108 mis-pointing is indicated. During installation, thediagnostics described also have value. Analyzing a combination ofmatched filter and AGC results, information to aid in the initial ODUantenna pointing and verification of the final installation can beprovided. The above examples, e.g., the use of state information from atleast one of the AGC, the ADC, the DSP, and the DAC providing adiagnostic output for the system, where the diagnostic output comprisesat least one of fault recognition, fault reporting, performancemonitoring, installation aiding, and installation verification for thesystem 100, are provided as examples of the class of diagnostics andinstallation verification aids possible by analysis of data related tothe state of the subsystems of the given architecture. Any suchdiagnostics or installation aids derived from such state information iswithin the scope of this invention.

Although discussed with respect to voltage control of DRO 804, othermethods of control of signals 806, and the effects of DRO 804 drift, canbe accomplished with the present invention. For example, and not by wayof limitation, a digital compensation within DSP 814 of the frequencyoffset, once estimated by drift estimator 822, can also be achieved viafeedback between drift estimator 822 and DSP 814.

CONCLUSION

The present invention discloses systems and devices for controllingfrequency drift in satellite broadcast systems. A receiver antennasystem for a direct broadcast satellite signal communications system inaccordance with one or more embodiments of the present inventioncomprises an oscillator, a mixer, coupled to the oscillator, forconverting satellite signals at a first frequency to signals at anintermediate frequency, an analog-to-digital (A/D) converter, coupled tothe mixer, for receiving the signals at the intermediate frequency andfor converting the signals at the intermediate frequency atnear-real-time to a digital data stream, a Digital Signal Processor(DSP), coupled to the A/D converter, for processing the digital datastream, and a drift estimator, coupled to the DSP, the drift estimatordetermining a frequency drift of the oscillator, wherein the receiverantenna system corrects the frequency drift of the oscillator using thedetermined frequency drift.

Such a system further optionally comprises the drift estimator driving adigital mixer within the DSP to compensate for the determined frequencydrift, an output of the drift estimator being fed back to the oscillatorto control the frequency drift of the oscillator, an automatic gaincontrol coupled between the mixer and the A/D converter, the A/Dconverter sampling the signals at the intermediate frequency at a speedgreater than 1 gigasample per second, the satellite signals beingtransmitted in at least the Ku-band of frequencies, the satellitesignals being further transmitted in at least the Ka-band offrequencies, a Digital-to Analog Converter (DAC), coupled to the DSP,and an output of the DAC being mixed with a second oscillator, such thatan output of the receiver antenna system is set to a desired band on asingle wire interface.

A system for distributing a plurality of satellite signals on a singleinterface in accordance with one or more embodiments of the presentinvention comprises an oscillator for down-converting the plurality ofsatellite signals to signals at an intermediate frequency, an AutomaticGain Controller (AGC) for gain controlling the signals at theintermediate frequency, an analog-to-digital (A/D) converter, coupled tothe AGC, for receiving the signals at the intermediate frequency,wherein the A/D converter directly samples the signals at theintermediate frequency, and a Digital Signal Processor (DSP), coupled tothe A/D converter, wherein a first output of the DSP is used todetermine the intermediate frequency and a second output of the DSP isan input to the single interface.

Such a system further optionally comprises a Digital-to-Analog Converter(DAC), coupled to the second output of the DSP, the first output of theDSP determining the intermediate frequency by controlling a frequency ofthe oscillator, the first output of the DSP driving a compensatoryfrequency shift by controlling a digital mixer internal to the DSP, theoscillator being a Dielectric Resonance Oscillator (DRO), the pluralityof satellite signals being transmitted in a plurality of frequencybands, the plurality of frequency bands comprising a Ka-band and aKu-band, the DRO down-converting the Ka-band to at least a firstintermediate frequency band and the DRO down-converting the Ku-band to asecond intermediate frequency band as convenient for A/D conversion, theA/D converter sampling the signals at the intermediate frequency at arate greater than the Nyquist rate for the signals at the intermediatefrequency, state information from at least one of the AGC, the ADC, theDSP, and the DAC providing a diagnostic output for the system, and thediagnostic output comprising at least one of fault recognition, faultreporting, performance monitoring, installation aiding, and installationverification for the system.

The foregoing description of the preferred embodiment of the inventionhas been presented for the purposes of illustration and description. Itis not intended to be exhaustive or to limit the invention to theprecise form disclosed. Many modifications and variations are possiblein light of the above teaching. It is intended that the scope of theinvention be limited not by this detailed description, but by the claimsappended hereto and the full range of equivalents of the claims appendedhereto.

1. A receiver antenna system for a direct broadcast satellite signalcommunications system, comprising: an oscillator; a mixer, coupled tothe oscillator, for converting satellite signals at a first frequency tosignals at an intermediate frequency; an analog-to-digital (A/D)converter, coupled to the mixer, for receiving the signals at theintermediate frequency and for converting the signals at theintermediate frequency at near-real-time to a digital data stream; aDigital Signal Processor (DSP), coupled to the A/D converter, forprocessing the digital data stream; and a drift estimator, coupled tothe DSP, the drift estimator determining a frequency drift of theoscillator and driving a digital mixer within the DSP to compensate forthe determined frequency drift, wherein the receiver antenna systemcorrects the frequency drift of the oscillator using the determinedfrequency drift.
 2. The receiver antenna system of claim 1, wherein anoutput of the drift estimator is fed back to the oscillator to controlthe frequency drift of the oscillator.
 3. The receiver antenna system ofclaim 1, further comprising an automatic gain control coupled betweenthe mixer and the A/D converter.
 4. The receiver antenna system of claim1, wherein the A/D converter samples the signals at the intermediatefrequency at a speed greater than 1 gigasample per second.
 5. Thereceiver antenna system of claim 1, wherein the satellite signals aretransmitted in at least the Ku-band of frequencies.
 6. The receiverantenna system of claim 5, wherein the satellite signals are furthertransmitted in at least the Ka-band of frequencies.
 7. The receiverantenna system of claim 6, further comprising a Digital-to AnalogConverter (DAC), coupled to the DSP.
 8. The receiver antenna system ofclaim 7, wherein an output of the DAC is mixed with a second oscillator,such that an output of the receiver antenna system is set to a desiredband on a single wire interface.
 9. A system for distributing aplurality of satellite signals on a single interface, comprising: anoscillator for down-converting the plurality of satellite signals tosignals at an intermediate frequency; an Automatic Gain Controller (AGC)for gain controlling the signals at the intermediate frequency; ananalog-to-digital (A/D) converter, coupled to the AGC, for receiving thesignals at the intermediate frequency, wherein the A/D converterdirectly samples the signals at the intermediate frequency; and aDigital Signal Processor (DSP), coupled to the A/D converter, wherein afirst output of the DSP determines the intermediate frequency and drivesa compensatory frequency shift by controlling a digital mixer internalto the DSP and a second output of the DSP is an input to the singleinterface.
 10. The system of claim 9, further comprising aDigital-to-Analog Converter (DAC), coupled to the second output of theDSP.
 11. The system of claim 9, wherein the first output of the DSPdetermines the intermediate frequency by controlling a frequency of theoscillator.
 12. The system of claim 11, wherein the oscillator is aDielectric Resonance Oscillator (DRO).
 13. The system of claim 9,wherein the plurality of satellite signals are transmitted in aplurality of frequency bands.
 14. The system of claim 13, wherein theplurality of frequency bands comprises a Ka-band and a Ku-band.
 15. Thesystem of claim 14, wherein the DRO down-converts the Ka-band to atleast a first intermediate frequency band and the DRO down-converts theKu-band to a second intermediate frequency band as convenient for A/Dconversion.
 16. The system of claim 9, wherein the A/D converter samplesthe signals at the intermediate frequency at a rate greater than theNyquist rate for the signals at the intermediate frequency.
 17. Thesystem of claim 10, wherein state information from at least one of theAGC, the ADC, the DSP, and the DAC provide a diagnostic output for thesystem.
 18. The system of claim 17, wherein the diagnostic outputcomprises at least one of fault recognition, fault reporting,performance monitoring, installation aiding, and installationverification for the system.